The table describes the four possible ways to WIRE a feedback network onto an amplifier. Every feedback amplifier has to do two jobs, and each job has exactly two options; the four topologies are just the four combinations. Once you see the two jobs, every word in the table unpacks itself.

Job 1 at the output: “samples” = what the feedback network measures

The feedback network must measure the output to know what to correct. There are only two things it can measure, and each forces a wiring style:

Sample the VOLTAGE: connect the network ACROSS the output, in parallel, exactly like a voltmeter. A parallel connection is called SHUNT. Sample the CURRENT: break the output loop and insert the network IN the current’s path, exactly like an ammeter. That is a SERIES connection.

Job 2 at the input: “mixes” = how the returned signal is subtracted

The returned correction must be combined with (subtracted from) the incoming signal. Two ways again: series mixing puts the feedback voltage into the same loop as the input source, so the VOLTAGES subtract around the loop (KVL does the subtraction). Shunt mixing joins the feedback wire and the input wire at one NODE, so the CURRENTS subtract at the junction (KCL does the subtraction).

The names: input word first, output word second

“SERIES-SHUNT” means: series mixing at the input, shunt (voltage) sampling at the output. “SHUNT-SERIES” is the mirror image: currents subtracted at the input node, output current sampled. The other two are the remaining combinations. (Watch out in other textbooks: some name the output first. This course’s convention is input-output.)

Z_in raised/lowered: feedback pushes impedances toward the ideal

The pattern behind those columns is one sentence: negative feedback makes the amplifier look MORE IDEAL for the job its topology defines, each by the familiar factor $(1+A\beta )$.

At the input: series mixing RAISES $Z_{\mathrm{in}}$ (good, a voltage input wants to draw no current), shunt mixing LOWERS it (good, a current input wants to present no obstacle). At the output: voltage sampling LOWERS $Z_{\mathrm{out}}$ (a stiff voltage source that doesn’t sag under load), current sampling RAISES it (a stiff current source that pushes its current regardless of load).

The “stabilises” column: four gains, two of them with odd names

Each topology holds one gain constant, the ratio of (what it samples) to (what its input naturally accepts):

Voltage gain $A_v=V_{\mathrm{out}}/V_{\mathrm{in}}$ (dimensionless). Current gain $A_i=I_{\mathrm{out}}/I_{\mathrm{in}}$ (dimensionless). Transconductance $G_m=I_{\mathrm{out}}/V_{\mathrm{in}}$: voltage in, current out, units A/V (siemens). Transresistance $R_m=V_{\mathrm{out}}/I_{\mathrm{in}}$: current in, voltage out, units V/A (ohms). The prefix TRANS- just says the ratio crosses from input to output with DIFFERENT quantities on top and bottom, so the “gain” carries units.

Familiar faces: the op-amp non-inverting amplifier is series-shunt (input voltage subtracted in a loop against the divider’s feedback voltage; output voltage sampled by the R₁-R_F divider), which is exactly why it has huge $Z_{\mathrm{in}}$ and tiny $Z_{\mathrm{out}}$. The inverting amplifier behaves as shunt-shunt (currents subtract at the virtual-earth node; output voltage sampled), which is why its input impedance is only $R_1$ rather than huge.

One-liner to keep: first word = how the input subtracts (series = voltages in a loop, shunt = currents at a node); second word = what the output measurement wire looks like (shunt = voltmeter → voltage, series = ammeter → current); and feedback drives every impedance toward the ideal for that combination.

They are the SAME quantity, the amplifier’s upper cutoff frequency, measured in two different situations. The subscripts are doing all the work:

fₖ: the cutoff of the bare (open-loop) amplifier

The subscript H stands for HIGH: it marks the high-frequency corner, the frequency (in Hz) where the amplifier’s gain has fallen 3 dB below its flat low-frequency value and begins rolling off. Up to $f_H$ the bare amplifier gives you its full gain $A_0$; beyond it the gain slides away. So $f_H$ is the bandwidth edge of the amplifier WITHOUT any feedback applied.

fₖₗ: the cutoff of the same amplifier WITH feedback

The extra subscript f stands for “with Feedback”, exactly the same convention as $A_f$ (closed-loop gain) versus $A$ (open-loop gain). Close the negative-feedback loop and the corner MOVES UP:

$$f_{\mathrm{Hf}}=f_H(1+A_0\beta )$$

Same amplifier, same physics; the feedback trades away gain (divides it by $1+A_0\beta$) and buys bandwidth with it (multiplies the corner by the SAME factor).

Why the corner moves at all

Intuition, not just algebra: near $f_H$ the bare amplifier’s gain starts to sag. But the feedback loop’s whole job is to hold the output at $v_s/\beta$ by amplifying the error harder whenever the output droops. So as the raw gain sags, the loop compensates, spending its surplus gain to keep the CLOSED-loop gain flat. It can keep this up until the raw gain has sagged all the way down to the closed-loop level, and that happens precisely at $f_{\mathrm{Hf}}$. The visible flat band stretches; the underlying roll-off never moved.

Both corners live on the same falling line, which is why the trade is exact:

$$A_{f0}·f_{\mathrm{Hf}}=A_0·f_H=\text{GBW (constant)}$$

One-liner to keep: fₖ = where the bare amplifier’s gain corner sits; fₖₗ = where that corner sits after feedback, pushed up by exactly the factor (1+A₀β) that the gain gave away. Subscript H = high-frequency corner; trailing subscript f = with feedback.

No, but the confusion is understandable, because the active high-pass filter in this course is BUILT FROM the inverting op-amp. The relationship is template and specialisation: the inverting op-amp is a general machine, and the filters are what you get when you fill in its two slots with particular components.

The template: gain = −Z₂/Z₁

The inverting amplifier is defined by its wiring: input through an impedance $Z_1$ into the − input, feedback impedance $Z_2$ from output back to the − input, + input grounded. Its gain is always

$$A_v=-\frac{Z_2}{Z_1}$$

The template says nothing about WHAT $Z_1$ and $Z_2$ are. That choice is where the personality comes from.

How each specialisation works

Plain amplifier: both slots are resistors. Resistors don’t care about frequency, so the gain $-R_2/R_1$ is flat: every frequency is amplified equally. An amplifier, not a filter.

Active low-pass: put a capacitor ACROSS the feedback resistor. At low frequency the capacitor is effectively open and invisible, so gain is $-R_2/R_1$; at high frequency its impedance collapses toward zero, it short-circuits $R_2$, so $Z_2\to 0$ and the gain dies. Highs killed = low-pass.

Active high-pass: put a capacitor IN SERIES with the input resistor. At low frequency the capacitor’s impedance is enormous, so $Z_1$ is huge and the gain $-Z_2/Z_1$ collapses: nothing gets in. At high frequency the capacitor is nearly a short, leaving $Z_1\approx R_1$ and full gain $-R_2/R_1$. Lows killed = high-pass.

Why the confusion is natural, and the sentence to keep

Every one of these circuits LOOKS like an inverting op-amp, because every one of them IS one; and a question about any of them starts from the same master formula $-Z_2/Z_1$. Also note the high-pass is not the only filter wearing this costume, and the inverting amp is not the only high-pass: the PASSIVE high-pass (just C then R, no op-amp) filters the same shape with no gain and no inversion.

So: “inverting op-amp” names a circuit TEMPLATE; “high-pass filter” names a BEHAVIOUR. The active high-pass is the template wearing a series input capacitor; with the capacitor moved to the feedback it becomes a low-pass; with plain resistors it’s no filter at all.

This formula answers one question: when a sine wave passes through a high-pass filter, by how much is the output shifted in time relative to the input? Piece by piece:

φₖₚ (phi): the phase shift of the output

${\phi }_{\mathrm{HP}}$ is an ANGLE, measured in degrees (or radians), that describes how far the output waveform is slid along the time axis compared to the input. One full cycle of the sine is 360°, so a shift of 90° means the output is displaced by a quarter of a cycle. The subscript HP just says this is the high-pass filter’s version of the formula (the low-pass has its own, with a minus sign and the fraction flipped).

The + sign is information, not decoration: positive phase means the output LEADS the input (its peaks happen earlier in time). The high-pass filter’s capacitor-first structure makes the output current, and hence the output voltage across R, peak before the input voltage does. A low-pass does the opposite: its output LAGS (negative φ), because the capacitor needs time to charge.

f: the frequency of the signal you are feeding in

$f$ is the frequency, in Hz, of the sine wave currently going through the filter. It is the variable you plug in: ask “what does the filter do at 200 Hz?” and 200 Hz is $f$. Different frequencies get different phase shifts, which is the whole point of the formula.

fₔ: the filter’s cutoff frequency

$f_c$ is the corner/cutoff frequency of the filter itself, fixed by the components: $f_c=1/(2\pi RC)$. It is a property of the circuit, not of the signal. In the formula it acts as the yardstick the signal frequency is compared against: only the RATIO $f_c/f$ matters.

arctan: the angle-recovery function

$arctan$ (inverse tangent, the tan⁻¹ button on your calculator) converts a ratio into the angle whose tangent equals that ratio. It appears because $R$ and the capacitor’s reactance $X_C$ sit at right angles in the complex plane; the phase angle of the divider is the angle of that right triangle, and “angle from a ratio of two perpendicular sides” is exactly what arctan computes. Make sure the calculator is in DEGREE mode if you want degrees out.

Reading the formula’s behaviour

Sanity anchors to memorise with it, all straight from the formula:

Far below cutoff ($f\ll f_c$): the ratio $f_c/f$ is huge, and arctan of a huge number approaches 90°: the output leads by nearly a quarter cycle (and is also tiny, since this is the stop band).

Exactly at cutoff ($f=f_c$): the ratio is 1 and $arctan(1)=45°$. The corner is always the 45° point (and the 0.707 amplitude point, both at once).

Far above cutoff ($f\gg f_c$): the ratio shrinks to zero, arctan(0) = 0°: in its pass band the high-pass hands the signal through unshifted and at full size.

Worked micro-example: high-pass with $f_c=1\text{ kHz}$, signal at $f=500\text{ Hz}$. Ratio $=1000/500=2$; $arctan(2)\approx 63°$. Output leads by about 63° (and is attenuated, since 500 Hz is below the corner).

An inductor is a coil of wire, and it does one thing: it gives current inertia. The current flowing through the coil builds a magnetic field around it; that field stores energy, and the stored energy resists any attempt to change the current, in either direction. A capacitor is a bucket for CHARGE whose voltage resists change; an inductor is a flywheel for CURRENT whose current resists change. Almost everything inductors are used for is this one property wearing different clothes.

The defining law, read slowly

$$v=L\frac{di}{dt}$$

The voltage across an inductor depends not on the current but on how fast the current is CHANGING. $L$ (henries, H) is how much inertia the coil has. Three consequences fall straight out, and every one of them is exam material:

1. Inductor current cannot jump. A step change in current would need $di/dt\to \infty$, hence infinite voltage. So current through an inductor is always a smooth, continuous ramp. (The exact dual of the capacitor, whose VOLTAGE cannot jump.)

2. To steady current it is a wire; to fast wiggles it is a wall. If $i$ is constant, $di/dt=0$ so $v=0$: at DC an ideal inductor is a short circuit. For AC its impedance is $Z_L=j\omega L$, growing with frequency: the higher the frequency, the harder it pushes back. Again the mirror image of the capacitor (which blocks DC and passes high frequency). That is why a series inductor used as a “choke” lets the DC through and chokes the ripple.

3. Interrupt the current and it bites. Open a switch feeding an inductor and you force a huge negative $di/dt$; the coil answers with a huge voltage spike of whatever polarity keeps its current flowing, arcing across the switch if necessary. This is why every relay coil and every converter has a freewheeling (flyback) diode: a path for the inductor current to continue into while it winds down gently.

The energy view (why PE2 is built on them)

The magnetic field holds energy

$$E=\frac{1}{2}L{I}^2$$

and unlike a resistor, an ideal inductor never wastes any of it; it only borrows and returns. That makes it the working component of every switching converter: close the switch and the inductor charges up (current ramps, energy in); open the switch and it insists on keeping its current flowing, pushing that energy on into the load through the diode. Do this tens of thousands of times a second and you move power from one voltage to another with almost no loss. Buck, boost and buck-boost converters differ only in WHERE the inductor sits and hence what the release path does; the duty-cycle formulas all come from the inductor ramp equation $\Delta i=(V/L)\Delta t$ balancing over one cycle.

Where you meet them

PE2: the converter inductor (energy shuttle), freewheeling diodes protecting switches, chokes smoothing rectifier output current, and transformers/motors, which are inductors with company (two coupled coils; coils on a rotating machine). AE2: LC tuned circuits and higher-order filters (though AE2 mostly builds filters from RC + op-amps instead). EM2: where $L$ itself comes from, computed from geometry via flux linkage, $L=N\Phi /I$.

One-liner to keep: an inductor stores energy in its magnetic field and uses it to keep its current smooth and continuous: a short to DC, a wall to fast change, a spike generator when interrupted, and an energy shuttle when switched on purpose.

Mains hum is the 50 Hz signal from the power grid leaking into circuits where it doesn’t belong. The UK grid alternates at 50 Hz (60 Hz in the US), and every mains cable, transformer and appliance around you is effectively a weak 50 Hz transmitter. If any of that couples into an audio or sensor circuit, you hear or measure a continuous low buzz at 50 Hz and its harmonics (100 Hz, 150 Hz…). That buzz is the “hum”: the deep drone you hear from cheap speakers, guitar amps with poor cables, or any circuit with its input lead dangling.

How it gets in (three doors)

1. Capacitive (electric-field) pickup. A mains wire and your signal wire form a tiny capacitor through the air. Tiny capacitance, but a high-impedance input needs almost no current to develop millivolts across it. This is why touching a scope probe with your finger displays a 50 Hz wave: your body is the antenna.

2. Magnetic (inductive) pickup. Mains current creates an alternating magnetic field; any LOOP of wire in your circuit is a one-turn transformer secondary in that field, and an EMF at 50 Hz is induced around it. Bigger loop area = more hum, which is why signal and return wires are kept together (twisted pairs make the loop area almost zero and alternate its sign every twist).

3. Power-supply ripple. Your own DC supply is made FROM the mains by rectifying and smoothing it. Imperfect smoothing leaves a residue riding on the DC rail; after a full-wave bridge rectifier the residue is at 100 Hz (both half-cycles get folded up, so the ripple frequency is twice the mains). If the rails wobble, the wobble leaks into the output. This is exactly the ripple calculation in the rectifier exam question: a bigger smoothing capacitor means smaller ripple, less hum.

Why AE2 cares: small signals lose

A sensor signal can be a few millivolts. If the wiring picks up even 10 mV of hum, the interference is bigger than the information, and an amplifier is loyal to neither: it amplifies both by the same gain. Filtering is awkward here because 50 Hz is often INSIDE the band you want (audio, ECG). The real fix exploits geometry instead:

Run the signal on TWO wires and subtract them at a differential amplifier. The wanted signal is applied BETWEEN the wires (differential), but hum couples onto both wires almost identically (common mode) because they sit side by side in the same field. Subtraction keeps the difference and cancels what is common: signal survives, hum vanishes. How well a real amplifier pulls this off is its CMRR (common-mode rejection ratio), and that is precisely why the instrumentation-amplifier exam question quotes a CMRR and asks for the smallest differential input you can still discern against interference.

The defence checklist

Keep loops small (twisted pair), shield cables (grounded screen intercepts capacitive pickup), avoid ground loops (two paths to ground = one big loop = one-turn secondary), smooth and regulate supplies (kills the 100 Hz ripple door), and measure differentially with high CMRR when the signal is small. Musicians’ folklore, “it stopped buzzing when I touched the metal chassis”, is this physics: your touch grounded the shield.

One-liner to keep: mains hum = 50 Hz leakage from the grid, entering by capacitance, induction, or supply ripple; it is common-mode on a wire pair, which is why differential amplifiers with high CMRR are the cure.

The slide is using one direction of a two-way pair. For VOLTAGE gain, the definitions are:

Ratio → dB (the definition):

$$\text{gain in dB}=20{\log }_{10}(A_v)$$

dB → ratio (the same equation solved for $A_v$, which is what the slide used):

$$A_v={10}^{\text{dB}/20}$$

To undo “20 log”, divide by 20 and raise 10 to it; the two operations are inverses, nothing more. With 40 dB: ${10}^{40/20}={10}^2=100$.

The power version (the 20 vs 10 trap)

For POWER ratios the factor is 10, not 20:

$$\text{dB}=10{\log }_{10}(P_2/P_1)\leftrightarrows \frac{P_2}{P_1}={10}^{\text{dB}/10}$$

The decibel is DEFINED on power with the 10. Voltage gets 20 only because power goes as voltage squared ($P=V^2/R$), and the square jumps out of the log as a factor of 2: $10{\log }_{10}{(A_v)}^2=20{\log }_{10}(A_v)$. Exam rule of thumb: voltages and currents use 20, powers use 10.

Landmarks that make the calculator optional

Because logs turn multiplication into addition, a handful of anchors covers most numbers you meet:

×10 = 20 dB, so ×100 = 40 dB, ×1000 = 60 dB (add 20 per factor of 10). ×2 ≈ 6 dB, ×√2 ≈ 3 dB, and ×1 = 0 dB (unity). Negative dB is attenuation: −20 dB = ÷10, −3 dB = ÷√2 ≈ 0.707, which is exactly why the bandwidth edge is called the −3 dB point: the voltage has fallen to 70.7%, i.e. HALF the power.

Combine by adding: 46 dB = 40 + 6 → 100 × 2 = 200. And 37 dB (the slide’s corner) = 40 − 3 → 100 ÷ √2 ≈ 70.7.

So the general machinery is just: dB = 20 log₁₀(ratio) for voltage (10 for power), and ratio = 10^(dB/20) to come back (10^(dB/10) for power). Everything else, cascades adding in dB, Bode plots being straight lines, the −3 dB bandwidth, is that one pair of formulas doing its multiplication-into-addition trick.

The two little arrows ($V^{+}$ pointing up, $V^{-}$ pointing down) ARE connections; they are just drawn in shorthand. An arrow like that means: “this pin goes off to the supply rail of that name, wherever it is drawn (or not drawn) on the page.” Typically $V^{+}=+15\text{ V}$ and $V^{-}=-15\text{ V}$ from a dual (split) supply. Nothing is floating; the wire to the batteries is simply not drawn, because if every diagram drew both supply wires to every chip, the signal path you are supposed to study would be buried in plumbing.

Why the op-amp must have these pins at all

An op-amp is an ACTIVE device. The energy in its output does not come from the input signal; a microvolt-level input could never light up a big output swing by itself. The output energy comes from the supply rails, and the input merely steers how much of it is delivered. Think of the input as the driver’s foot and the supply as the fuel tank: no tank, no motion, however hard you press. Remove the supply and the op-amp is an inert lump of silicon; every gain formula silently assumes the rails are there.

Two places the “invisible” rails still bite you in the exam

Saturation: the output physically cannot go above $V^{+}$ or below $V^{-}$; in practice it stops a volt or two inside them, at $\pm V_{\mathrm{sat}}$. Ask a gain-100 amplifier on ±15 V rails to amplify 0.5 V and it will NOT give 50 V; it slams into ≈±13–14 V and the waveform's tops get clipped flat. Comparator, Schmitt-trigger and oscillator questions run on exactly this: their output is only ever at one rail or the other.

Single vs split supply: with a split supply ($+15/-15$) the output can swing both sides of 0 V, which is why AC signals centred on ground work directly. Power a circuit from a single rail ($0$ to $+15$) and the output can never go negative; the signal has to be biased up to mid-rail instead.

Same convention elsewhere: ground symbols are the identical trick in the other direction. Every little ground pointing down means “connected to the common 0 V node”, and all of them are one node even though no wire is drawn between them. Circuit diagrams omit the boring, always-the-same wiring (power in, common return) so the interesting wiring stands out; the arrows and ground stubs are the breadcrumbs it leaves behind.

Short answer: within the region your exam questions live in, yes, and that is deliberate; but it is not a law of nature. −20 dB/decade is the fingerprint of exactly ONE pole (one RC time constant) being in charge. Wherever one pole dominates, you get exactly that slope; wherever more poles join in, the slope steepens in −20 steps.

Why one pole gives exactly −20 and nothing else

A single pole means the gain behaves like

$$|A|=\frac{A_0}{\sqrt{1+{(f/f_c)}^2}}$$

Well above the corner ($f\gg f_c$) the 1 under the root stops mattering and this collapses to $|A|\approx A_0{f}_c/f$: gain is proportional to $1/f$. Multiply frequency by 10 and the gain divides by exactly 10, which IS −20 dB. The slope cannot be anything else, and it cannot drift: it stays exactly −20 dB/decade for as long as that one pole is the only thing acting. That perfect consistency is precisely why GBW = constant works.

Where it stops being true

Three boundaries, all visible on the sketch:

Below the corner $f_c$: the gain is flat, slope 0. No roll-off yet.

Between the corner and (nearly) $f_T$: the −20 dB/decade region. In a general-purpose op-amp this stretch is enormous, maybe 10 Hz to 1 MHz, five decades, and it is CONSISTENT the whole way because manufacturers build in a compensation capacitor that creates one deliberately dominant pole far below all the others. The op-amp is engineered to look like a textbook single-pole system over its whole useful range. This is the only region your GBW exam questions use.

Near and beyond $f_T$: the transistors’ own parasitic poles catch up. Each extra pole adds another −20, so the plot bends to −40, then −60 dB/decade. (This steepening region is also where feedback can turn unstable, which is exactly WHY the makers force the single dominant pole below it; that story belongs to control theory, not the AE2 exam.)

The rule as a slope bookkeeping system

The generalisation worth keeping: every pole contributes −20 dB/decade, every zero contributes +20, and slopes simply add. You already use this in the filters unit: a first-order RC filter rolls off at −20 dB/decade; a second-order filter at −40; an n-th order at $n\times (-20)$. When an exam asks you to sketch a Bode plot, you are just placing corners and adding ±20 slopes between them.

So: consistent, yes, wherever a single pole dominates, and op-amps are purpose-built so that this covers everything you will calculate. Just remember the slope is per-pole bookkeeping, not a universal constant: more poles, steeper line.

“Unity” is just the formal word for the number 1. Unity gain = a gain of exactly 1: the output is a copy of the input, same size, not amplified, not attenuated. In decibels that is

$$20{\log }_{10}(1)=0\text{ dB}$$

so “unity gain”, “gain of 1” and “0 dB” all say the same thing. The phrase shows up in three different corners of AE2, and each one is exam-relevant.

1. The unity-gain buffer (voltage follower): the amp that “does nothing” on purpose

Tie the op-amp’s output straight back to its inverting input and feed the signal into the + input:

Golden Rule 2 forces $V_{-}=V_{+}$, and here $V_{-}$ IS the output, so $V_{\mathrm{out}}=V_{\mathrm{in}}$. Same result from the non-inverting formula with all of the output fed back ($\beta =1$): $A_v=1+R_F/R_1$ with $R_F=0$ gives 1.

Why build an amplifier that doesn’t amplify? Because gain was never the point; muscle is. The follower has a nearly infinite input resistance (it barely disturbs the delicate circuit driving it) and a tiny output resistance (it can drive a heavy load without sagging). It is an impedance converter: same voltage, vastly better ability to deliver it. Classic use: a sensor that can only supply microamps needs to drive something hungry; put the follower in between and neither side troubles the other (this is the cure for the loading problem in the previous entry).

2. The unity-gain frequency: where an op-amp runs out of gain

A real op-amp’s open-loop gain is huge at DC but falls steadily with frequency. The frequency where it has dropped all the way to 1 (0 dB) is the unity-gain frequency $f_T$, and for the usual one-pole roll-off it equals the gain-bandwidth product:

$$\text{GBW}=\text{gain}\times \text{bandwidth}=f_T$$

This is the number the paper hands you (“GBW = 1 MHz”) so you can trade gain for bandwidth: build a ×100 amplifier from it and your bandwidth is $1\text{ MHz}/100=10\text{ kHz}$. The follower, using gain 1, gets the entire 1 MHz, another reason buffers are everywhere.

3. Cousins worth recognising

The transistor versions of the buffer, the emitter follower (BJT) and source follower (FET), have gain “approximately unity” for the same job: high input impedance, low output impedance, gain ≈ 1 (slightly under, e.g. 0.95, since they lack the op-amp’s feedback precision). And in the oscillator unit, the Barkhausen condition demands unity LOOP gain, $|A\beta |=1$: once around the loop the signal must come back at exactly its own size, so it neither dies out nor blows up.

So whenever you meet the phrase: unity gain = 1× = 0 dB, and the interesting question is never the number itself but WHY a gain of 1 is useful there: copying a voltage with muscle (follower), running out of amplification ($f_T$), or sustaining itself exactly (oscillator).

The load is whatever you connect to a circuit’s output to USE the signal: the speaker on an audio amp, the next amplifier stage, a motor, an ADC input, or in exam questions simply a resistor called $R_L$. From the circuit’s point of view, everything hanging on its output terminals behaves like one resistance that draws current, so we model the whole consumer as that single $R_L$.

Why it has a dramatic name and not just “R3”

Because connecting it COSTS the circuit something, the way loading cargo costs a lorry speed. The amplifier’s output isn’t a perfect voltage source; it has an internal output resistance $R_{\mathrm{out}}$ in series. The moment a load draws current, that current flows through $R_{\mathrm{out}}$ and drops voltage across it, so LESS of the internal voltage reaches the outside. The two resistances form a voltage divider:

$$v_L=K_v{v}_{\mathrm{in}}·\frac{R_L}{R_{\mathrm{out}}+R_L}$$

That reduction is called the loading effect, and “load” also works as a verb: “stage B loads stage A” means B’s input resistance drags A’s output down exactly this way. This is the core mechanic of every cascade question in the paper.

Heavy load vs light load (the counterintuitive bit)

A heavy load is a SMALL $R_L$: small resistance draws lots of current and drags the output far down (divider fraction small). A light load is a LARGE $R_L$: barely any current, barely any drop. The extreme light load is nothing connected at all, called no load / unloaded / open circuit, where the divider fraction is 1 and you get the full $K_v{v}_{\mathrm{in}}$ (that is exactly the condition for measuring $K_v$, two entries up). Battery version of the same physics: a 9 V battery reads 9 V on a voltmeter (light load) but sags to 7 V cranking a motor (heavy load).

Same word elsewhere in AE2

Load line (diode/BJT graph questions): the straight line $I=(V_s-V)/R_L$ expressing what the source-plus-load-resistor part of the circuit permits; the device’s curve crosses it at the operating point. Same $R_L$, same idea: the load constrains the device.

And the design takeaway the course keeps repeating: a good voltage amplifier wants $R_{\mathrm{out}}\ll R_L$ (and $R_{\mathrm{in}}$ huge so IT is a light load on whatever feeds it). Then the divider fractions are ≈1 and the gain barely notices what you connect.

Feedback means: take a sample of the output and send it BACK to the input, so the circuit can react to its own result. That is the whole word. “Feed” a bit of the output “back”. Everything else in the unit is about what happens once you do that, and it splits into two opposite worlds depending on the SIGN with which the returned sample is combined.

Everyday version first

You already use feedback constantly. Steering a car: you look at where the car actually IS (the output), compare it with where you want it (the input), and turn the wheel by the DIFFERENCE. You do not steer blind with a pre-planned sequence of wheel angles; you continuously correct using the result. A thermostat is the same: measure the room, compare with the set temperature, heat by the error. Both are negative feedback: the correction opposes the error, so the error shrinks.

The circuit version: three parts

Three parts, and each has one job:

The amplifier $A$ is powerful but crude: huge gain, but an inaccurate, temperature-dependent, device-to-device-varying number. The feedback network $\beta$ is the opposite: it does nothing clever, it is usually just a resistor divider returning a fixed fraction of the output, but a resistor ratio is precise and stable. The summing node compares: it subtracts the returned sample from the input, and what is left, $x_i=x_s-x_f$, is called the error. Only the error gets amplified.

Why subtracting is magic (negative feedback)

Suppose the output drifts a little too HIGH. Then the returned fraction $x_f$ is too high, so the error $x_s-x_f$ DROPS, so the amplifier drives the output back DOWN. Drift low and the same chain pushes it back up. The loop punishes every deviation of the output from $x_s/\beta$; it is self-correcting, like your steering. With a huge $A$ the loop only settles when the error is almost exactly zero, which forces

$$x_f\approx x_s$$

and since $x_f=\beta x_o$, the output must sit at $x_o\approx x_s/\beta$. The exact formula from the lectures says the same thing without the approximation:

$$A_f=\frac{A}{1+A\beta }\to \frac{1}{\beta }\text{ when }A\beta \gg 1$$

Read what that trade is: you GIVE UP most of the raw gain (from $A$ down to $1/\beta$) and in exchange the gain you keep is set by the trustworthy resistor ratio instead of the untrustworthy transistor. That one trade buys all four advertised benefits: stable gain, less distortion, wider bandwidth, and tamed input/output impedances.

This is also the secret behind the op-amp golden rule “$V_{-}=V_{+}$”: it is not a property of the op-amp alone. The op-amp is just an enormous $A$; it is the NEGATIVE FEEDBACK loop around it that drives the error between the two inputs to essentially zero. No feedback (or feedback to the wrong pin) and the rule is dead.

The other sign: positive feedback

Flip the summing node to ADD the sample instead of subtracting it and every deviation is now rewarded: output rises → returned sample rises → input to the amplifier rises → output rises harder. A runaway. That is useless for amplifying but exactly what you want for two other things the unit covers: circuits that SNAP decisively between two states (Schmitt trigger, astable), and oscillators, where the loop feeds the signal back to sustain itself. An oscillator holds itself going when one trip around the loop returns the signal at exactly its original size and phase, the Barkhausen condition:

$$A\beta =1\text{ (and }0°\text{ phase shift round the loop)}$$

So one sentence to keep: feedback = the circuit listening to its own output; subtract the sample and it self-corrects into a precise amplifier with gain $1/\beta$; add the sample and it self-reinforces into a switch or an oscillator. The quantity deciding everything is the loop gain $A\beta$: negative feedback wants it large, oscillation wants it exactly 1.

Because the diagram is using + and − for two completely different jobs, and only one of them is the “inverting / non-inverting” labelling you were thinking of.

Job 1: the +/− INSIDE the triangle = which input is which

On the op-amp symbol, the − printed at one input and the + printed at the other are names, not voltages. They say: this pin is the inverting input ($V_1$ in the lecture’s model), that pin is the non-inverting input ($V_2$). One symbol per pin, fixed forever. They tell you which input gets the minus sign in

$$V_{\mathrm{out}}=A(V_2-V_1)$$

Job 2: the little +/− PAIRS in the model = polarity marks of a measurement

A “voltage” is never a property of one point. It is always a difference between TWO points, so every voltage you draw needs two reference marks: a + on the point you measure, a − on the point you measure from. The pair just defines which direction counts as positive.

In the internal model, $V_1$ means “the voltage of the inverting terminal measured relative to ground”. So the diagram puts a + at the terminal and a − at the ground symbol below it: red voltmeter lead on the pin, black lead on ground. Same for $V_2$. Each input therefore carries a full +/− PAIR, because each is its own little two-point measurement. It is not saying the input “is both positive and negative”.

Reading the numbers

With the pair placed + at the pin and − at ground, “$V_1=3\text{ V}$” means the inverting pin sits 3 V ABOVE ground. If the pin happened to sit below ground you would not move the marks; the number itself just comes out negative, $V_1=-2\text{ V}$. The marks fix the convention; the sign of the value does the rest.

The tall arrow between the two pins in your snip is a THIRD measurement, taken pin-to-pin without involving ground at all:

$$V_{\mathrm{DIFF}}=V_2-V_1$$

and that difference is the only thing the ideal op-amp ever looks at: it appears across $Z_{\mathrm{in}}$ and gets multiplied by the huge gain $A$.

So the rule to carry forward: ± printed inside the triangle = which pin is which (inverting vs non-inverting); a ± pair drawn beside a labelled voltage = the two reference points of that measurement. Every voltage in every circuit diagram in this course carries such a pair, op-amp or not; it is only here that the two conventions sit close enough to collide.

“Wiggle” is the informal name for the small AC signal part of a voltage or current. In an amplifier, every voltage and current is a big steady DC level with a tiny signal moving up and down on top of it. The steady level is the bias (the Q-point). The moving part is the wiggle, and the wiggle IS your signal; it is the only part the amplifier is trying to amplify.

$$i_C(t)=\underset{\text{DC bias}}{I_C}+\underset{\text{the wiggle}}{i_c}$$

The notation carries the split: capital letter, capital subscript ($I_C$) is the DC value with no signal; lowercase letter, lowercase subscript ($i_c$) is the wiggle alone; lowercase letter with capital subscript ($i_C$) is the total, the sum of both.

Why the word exists: it buys you a linear circuit

A transistor is a NONLINEAR device: its input-output curve is bent (exponential for a BJT). Nonlinear circuits are hard. But zoom in on any smooth curve close enough and it looks like a straight line. If the signal only wiggles a small distance about the Q-point, the transistor only ever uses a tiny, effectively straight piece of its curve, so for the wiggle alone the amplifier behaves as a LINEAR circuit whose gain is the slope of the curve at the Q-point (that slope is where $g_m$ comes from).

How the course uses it

The word appears whenever the analysis splits in two, and the split is the whole method of the BJT and FET amplifier units:

DC analysis (signal off): kill the wiggle, keep the batteries. Find the Q-point ($I_C$, $V_{\mathrm{CE}}$) from the bias resistors. This decides WHERE on the curve the amplifier sits.

Small-signal (AC) analysis (only the wiggle): kill the DC instead. Constant supplies have zero wiggle, so they become AC ground; big coupling capacitors pass the wiggle freely, so they become shorts; the transistor becomes its small-signal model. This linear circuit answers “how does the circuit look TO the wiggle?” and gives the gain.

Because the zoomed-in curve is straight, the two answers superpose: total = bias + amplified wiggle, which is exactly the first equation again.

One catch worth remembering for exam comments: the wiggle must actually BE small. Drive the input hard and the signal reaches the bent parts of the curve (or hits saturation and cut-off at the ends), the straight-line approximation dies, and the output comes out distorted, a flattened, non-sine shape. That is also why a measured gain is taken with a deliberately small input (see the Kᴸ measurement entry above).

One sentence first, then the detail: you feed the amplifier a small sine wave, look at the input and the output on an oscilloscope with nothing connected to the output, and divide the two amplitudes:

$$K_v=\frac{v_{\mathrm{out(open)}}}{v_{\mathrm{in}}}$$

Every word of that sentence is doing a job. Here is why each one is there.

The bench setup

Step by step

1. Drive it with a small mid-band sine. Set the generator to a sine of a few tens of millivolts at a middle frequency like 1 kHz. Small, so the amplifier stays in its linear region; mid-band, so coupling capacitors (which kill gain at low frequency) and stray capacitance (which kills gain at high frequency) are both out of the picture and you are measuring the flat, maximum gain.

2. Put CH1 across the amplifier’s INPUT terminals. Not on the generator’s dial setting. The generator has its own internal resistance $R_s$, and some of its voltage is lost across it before reaching the amplifier (the input divider). $v_{\mathrm{in}}$ in the definition of $K_v$ is the voltage that actually appears at the amplifier’s input pins, so that is where the probe goes.

3. Put CH2 across the output with NO load connected. This is the crucial one. $K_v$ is defined as the open-circuit gain: the gain when no current is drawn, so nothing is dropped across $R_{\mathrm{out}}$. If you attach a load $R_L$, the output divider steals part of the internal voltage and you would measure the smaller loaded gain:

$$\frac{v_{\mathrm{out}}}{v_{\mathrm{in}}}=K_v·\frac{R_L}{R_{\mathrm{out}}+R_L}$$

A scope probe is allowed because its resistance (1 MΩ or 10 MΩ) is thousands of times bigger than a typical $R_{\mathrm{out}}$ of a few kΩ: the divider fraction is essentially 1, so the probe is “open circuit” as far as the amplifier can tell.

4. Check the output is a clean sine, then read both amplitudes. If the tops of the output wave are flattened, the amplifier is clipping against its supply rails and you are measuring the power supply, not the gain: turn the input DOWN until the shape is clean. Then read the same measure off both channels (both peak-to-peak, or both amplitude, or both RMS; it cancels in the ratio) and divide:

$$K_v=\frac{v_{\mathrm{out}}\left(\text{pk-pk}\right)}{v_{\mathrm{in}}\left(\text{pk-pk}\right)}$$

Volts divided by volts, so $K_v$ is a pure number. Quote it in decibels if asked:

$$K_v\left(\text{dB}\right)=20{\log }_{10}{K}_v$$

The same trick measures Rₒᵤᵗ and Rᵢₙ too

The classic lab follow-up, worth knowing because it uses the same setup. First record the open-circuit output $v_{\mathrm{out(open)}}$. Now connect a variable resistor as the load and turn it down until the output reads exactly HALF of that. At that point the divider is $\frac{R_L}{R_{\mathrm{out}}+R_L}=\frac{1}{2}$, which forces $R_L=R_{\mathrm{out}}$: read $R_{\mathrm{out}}$ straight off the dial. Mirror image at the input: put a known resistor in series with the generator and increase it until $v_{\mathrm{in}}$ at the amplifier terminals halves; that resistor now equals $R_{\mathrm{in}}$.

So when a question says an amplifier “has $K_v=100$”, that number was obtained exactly this way: small clean sine in, unloaded sine out, ratio of the two. Everything else in the systems model (the loading formula with the two dividers) exists to predict how much LESS than $K_v$ you get once a real source and a real load are attached.

Two things that look alike but are different kinds of object: $K_v$ is a fixed NUMBER describing the amplifier; $K_v{v}_{\mathrm{in}}$ is that number MULTIPLIED by the input voltage, and it is the value of the internal voltage source in the amplifier’s model. Read the dot that isn’t written: Kᴸvᵢₙ means $K_v\times v_{\mathrm{in}}$.

Kᴸ: one number printed on the box

$K_v$ is the amplifier’s open-circuit (unloaded) voltage gain: how many times bigger the output is than the input when NOTHING is connected to drag it down. If $K_v=100$, the amp is a “multiply by 100” machine in the ideal case. It has no units (volts per volt) and it does not change with the signal; it is a property of the amplifier itself, like a car’s top speed.

The three-part model where Kᴸvᵢₙ lives

Circuit theory replaces the whole complicated amplifier with three parts: an input resistance $R_{\mathrm{in}}$ (what the source sees), a dependent voltage source whose value is $K_v{v}_{\mathrm{in}}$, and an output resistance $R_{\mathrm{out}}$ in series with it:

So $K_v{v}_{\mathrm{in}}$ is not a new symbol to memorise; it is a recipe: measure the voltage at the input terminals, multiply by $K_v$, and that is the voltage the internal source generates. If $v_{\mathrm{in}}=10$ mV and $K_v=100$, the source pumps $1$ V.

Why the distinction earns marks: real gain is LESS than Kᴸ

The exam trick is that $K_v$ only equals the measured gain when nothing loads the amplifier. In a real circuit two voltage dividers eat into it: at the input, the source’s own resistance $R_s$ and $R_{\mathrm{in}}$ split the signal before it is even measured; at the output, $R_{\mathrm{out}}$ and the load $R_L$ split the pumped voltage before it reaches the load:

That is exactly the 2022 Q1(b) pattern: an amplifier with $K_v=100$ (40 dB unloaded) delivers an overall gain of only $96.6$ (39.7 dB) once the dividers take their cut. The unloaded number describes the box; the loaded number describes the circuit you built with it.

The three-line summary

• 1$K_v$: a fixed, unitless NUMBER, the amplifier’s gain with no load (open-circuit gain).
• 2$K_v{v}_{\mathrm{in}}$: a VOLTAGE, the internal dependent source’s value: Kᴸ times whatever vᵢₙ currently is.
• 3Measured gain < $K_v$ in any real circuit, because the input and output dividers each take a slice. Quoting Kᴸ as the circuit gain is the classic lost mark.

The exams: ENGM2 Mon 27 Jul · EM2 Tue 28 Jul · AE2 Thu 30 Jul · PE2 Wed 5 Aug · EP2 Fri 7 Aug · MS2A Mon 10 Aug (all 09:30). Target: a solid 50% in each. The front is brutal (100 of the 155 hours land before 30 Jul, about 8h/day); the back half is comfortable (4–5h/day).

Phase 1, the crunch: 18–29 Jul (~8h/day)

• 1Every morning 18–29 Jul: the 2h AE2 session (list below), fresh brain, before anything else. Its value is the SPACING; missing sessions cannot be made up by cramming.
• 218–21 Jul, ~6h/day: ENGM2 (~26h). Fourier (ride the momentum), then Laplace (most formulaic), then first-order ODEs. Ruthlessly skip the MVC tail (Stokes/divergence proofs) and second-order ODE depth.
• 322–25 Jul, ~6h/day: EM2 (~26h). Vikram chapters + the eight recycled paper patterns ONLY: Gauss sphere/coax, B-fields of wires, induction/emf, capacitor energy, dielectric sphere, Maxwell statements. No derivation completionism.
• 426 Jul: final ENGM2 drill (papers, timed). 27 Jul after the exam: final EM2 drill. 28 Jul afternoon + 29 Jul: AE2 finale (mixed tutorial runs under time).

The 12 AE2 morning sessions (2h each)

• 1Op-amp golden rules + ideal properties (op-amp unit, first sections)
• 2Inverting amplifier: derivation, then gain practice until automatic
• 3Non-inverting amplifier + voltage follower/buffer
• 4Summing and difference amplifiers
• 5Decibels unit: landmarks table + log rules
• 6The no-calculator dB conversions + cascaded stages
• 7Gain-bandwidth product + frequency-response basics
• 8Active filters: first-order low-pass and high-pass
• 9Filter practice + response sketches
• 10Op-amp non-idealities (offset, slew) light pass + tutorial questions
• 11Mixed tutorial run, timed
• 12Mixed past-paper op-amp/dB questions, exam conditions

Deliberately absent: BJT/FET small-signal depth. That is >50% material and it is the sacrifice that makes this schedule possible.

Phase 2, the recovery: 31 Jul–9 Aug (4–5h/day)

• 531 Jul–4 Aug: PE2 (~15h) + start EP2 alongside. PE2 bundle: waveform RMS, rectification with smoothing, transformers, one converter, one thermal chain.
• 65 Aug afternoon + 6 Aug: finish EP2 (~15h total). Bundle: number systems, IEEE-754 encode/decode, ALU traces, memory dumps, reading GPIO C. Pure pattern drilling.
• 77 Aug afternoon + 8–9 Aug: MS2A (~25h). Axial/torsion/bending stress + Mohr’s circle from solved examples until mechanical.

The three rules that decide it

• !The next ten days are FULL-TIME. One lost day before 27 Jul has nowhere to be absorbed.
• !EM2 must not leak backwards into ENGM2’s four days; the maths feeds everything and goes first.
• !The daily AE2 block happens even on bad days. Spacing is the whole strategy for that subject.

The task, from the general recipe (with $\omega =1$ and the integral already shrunk to where the wave is 1):

Step 1: the antiderivative, checked, not assumed

We need a function whose derivative is $\sin (nt)$. The candidate is $-\cos (nt)/n$. Check by differentiating it:

Chain rule pushes out a factor $n$ and a minus; they cancel against the built-in $-1/n$, leaving exactly $\sin (nt)$. So both the minus and the divide-by-$n$ are compulsory.

Step 2: substitute the limits

The bracket notation means “the antiderivative at the TOP limit, minus it at the BOTTOM limit”. Top is $t=\pi$, bottom is $t=0$:

Step 3: subtracting a negative becomes adding

The second bracket was minus a negative, so it flips to plus. This is where most sign errors are born.

Step 4: cos(0) = 1

Zero angle means no turn at all, and cosine of no turn is 1 (the very top of the trig-values table):

Step 5: combine over the shared denominator

Both terms sit over $n$, so they merge into one fraction, written with the positive term first:

Step 6: bring back the prefactor

The $1/\pi$ has been waiting outside the bracket the whole time. Multiplying two fractions multiplies their denominators:

Step 7: split by even and odd n

Now use $\cos (n\pi )={(-1)}^n$ (its own entry below): the numerator is $1-1=0$ for even $n$, and $1-(-1)=2$ for odd $n$:

Concrete values, to make the pattern real:

• 1$b_1=\frac{2}{\pi }$  (odd: numerator 2)
• 2$b_2=0$  (even: numerator 0, the harmonic is simply absent)
• 3$b_3=\frac{2}{3\pi }$  (odd again, and smaller: higher harmonics contribute less)

That is every step between the integral and the answer. The same seven-step shape runs every bₙ and aₙ in this course; only the wave’s constants and limits change.

First, your real question: is there a general formula?

Yes. Every “find the Fourier series” question, this one, Q2(b), the exam’s, is solved by the SAME four lines. Nothing about them changes between questions; you only swap in your wave’s brace and period:

Read the template in words: any repeating wave = a constant + a mix of sines and cosines at frequencies 1×, 2×, 3×… the fundamental. The three integrals (the “Euler formulae”) measure HOW MUCH of each ingredient your particular wave contains. ${\int }_T$ means “integrate across any one full period”. The recipe order is always: find $T$ and $\omega =2\pi /T$ first, then the three integrals, then assemble.

What each symbol is, for a beginner

• ∑$\sum$ means “add up”: the terms for $n=1,2,3,\dots$ one after another, forever.
• n$n$ is the harmonic number: which ingredient we are on. $n=1$ wiggles once per period, $n=3$ three times, etc.
• a,b$a_n,b_n$ are plain NUMBERS (one per $n$): the amount of the $n$-th cosine and sine in the mix. Finding them IS the whole question.
• ∫An integral is an AREA machine: $\int fdt$ totals the area under the curve. Multiplying $f$ by $\cos$ or $\sin$ before integrating makes it measure the OVERLAP between your wave and that ingredient (the deep reason is orthogonality, covered in the Fourier unit; for solving questions you only need to run the machine).

Now run the recipe on Q2(a), every move shown

Ingredients: $h=1$ on $(0,\pi )$, $0$ on $(\pi ,2\pi )$. Window length $T=2\pi$, so $\omega =2\pi /2\pi =1$, which is why every trig argument below is just $nt$.

Coefficient 1: a₀ (four small moves)

• 1Prefactor: $2/T=2/2\pi =1/\pi$. Just arithmetic.
• 2Limits shrink from $(0,2\pi )$ to $(0,\pi )$ because $h=0$ on the second half, and integrating zero adds zero. The brace’s value (1) replaces $h$ on the part that remains.
• 3$\int 1dt$ from 0 to π = area of a rectangle, height 1, width π = $\pi$. No calculus needed once you see the rectangle.
• 4The series uses $a_0/2=1/2$, the wave’s average height (the DC term entry below).

Coefficient 2: aₙ (why every cosine dies)

Moves: same prefactor and limit-shrink as before. The antiderivative of $\cos (nt)$ is $\sin (nt)/n$, divide by $n$ because differentiating $\sin (nt)$ spits out a factor $n$ (chain rule) that must be cancelled. Evaluate top minus bottom: $\sin (n\pi )$ is the sine of a whole number of half-turns, ALWAYS 0 (the trig-values table entry), and $\sin (0)=0$. So every single cosine coefficient is zero, which the hidden-symmetry argument predicted without integrating.

Coefficient 3: bₙ (the one that survives)

Two sign traps: the antiderivative of $\sin$ carries a built-in MINUS, and evaluating the bracket top-minus-bottom turns that into $1-\cos (n\pi )$. Every micro-move of that evaluation is worked one line at a time in the entry above this one. Then the even/odd split (the $\cos (n\pi )={(-1)}^n$ entry): even $n$ gives $1-1=0$; odd $n$ gives $1-(-1)=2$:

Assemble: pour the numbers back into the template

The template wants $a_0/2$ plus all the surviving terms. $a_0/2=1/2$; every $a_n$ is 0 (no cosines); $b_n$ is 0 for even $n$ and $2/n\pi$ for odd $n$, so only $n=1,3,5,\dots$ appear:

Sanity-read the answer: the ½ is the average height (right, the wave spends half its time at 1). Only sines, because wave-minus-mean is odd. Only odd harmonics, coefficients shrinking like 1/n because the wave has jumps. Every feature of the answer was predictable from the graph, which is the real power move in the exam.

The takeaway

• 1ONE general recipe solves every such question: $\omega =2\pi /T$, then the three Euler integrals, then assemble into the template. Memorise those four lines.
• 2What changes per question is only the BOOKKEEPING: your wave’s brace sets the integral limits and the constants inside.
• 3The recurring micro-skills: antiderivatives of $\cos /\sin (nt)$ (divide by $n$, sine’s carries a minus), $\sin (n\pi )=0$, $\cos (n\pi )={(-1)}^n$. Those three facts finish almost every coefficient in this course.

For reading the point ITSELF: correct, the middle of a flat piece always has a clean, unambiguous height, as far from every jump as possible. But there are two ways “pick the middle” can still bite, and one of them happens in your own Q2(b).

Caveat 1: your point is safe, but its MIRROR PARTNER might not be

The symmetry test uses two points: $t$ and $-t$. Choosing $t$ mid-shelf guarantees nothing about where $-t$ lands. Concrete case, the Q2(b) wave: $h=-1$ on $(-2,0)$, $+1$ on $(0,1)$, period 3, jumps at …, −2, 0, 1, 3, …

• ✗Middle of the WIDE shelf: $t=-1$. Perfectly readable ($h=-1$). But its partner is $+1$, which is EXACTLY the seam where the wave jumps from +1 down to −1. Partner unreadable, test dead on arrival.
• ✓Middle of the NARROW shelf: $t=0.5$. Partner $-0.5$ sits comfortably inside the wide shelf. Both readable: $h(0.5)=+1$, $h(-0.5)=-1$. Test works.

So the complete habit is: middle of a shelf for your point, then CHECK where the partner lands. If it hits a seam, slide your test time a little (0.4 instead of 0.5, say) or pick the other shelf, any interior pair works.

Caveat 2: a middle point that PASSES proves nothing

Safety of reading is not the same as strength of conclusion. If your middle-point test FAILS, done, the symmetry is disproved forever. If it PASSES, you have learned almost nothing, symmetry must hold at EVERY point, and one agreeing pair cannot certify that. Q2(b) again: points near the origin pass the odd test (heights flip nicely), yet the wave is not odd, which a second pair (like $t=1.5$) exposes. When a test passes, either test more pairs, or argue from the whole shape (equal widths? never negative? mirror image by construction?).

The complete checklist

• 1Pick $t$ strictly inside a shelf (middle = maximum safety margin). Never $0$, never a jump.
• 2Locate $-t$ (use periodicity if needed) and confirm IT is inside a shelf too. If not, adjust $t$.
• 3A FAIL is final; a PASS demands more evidence (another pair, or a whole-shape argument).

Short answer: the brace IS the formula. There is no single algebraic expression like $h(t)=t^2$ hiding somewhere; the piecewise rule is the complete definition, and “plugging in” works differently for it.

Two kinds of function definitions

• 1Formula-functions, like $f(t)=t^2$: plugging in means COMPUTING. Feed in 3, do the arithmetic, get 9.
• 2Rule-functions (piecewise), like our $h$: plugging in means LOOKING UP. Feed in a time, check which case it falls into, read off that case’s value. No arithmetic on the value at all.

Both are complete, legitimate definitions. Maths does not require a function to be one algebraic expression; it only requires that every input gets exactly one output. The brace does that job by cases.

The FULL definition of h has two parts

The brace covers one window, $0<t<2\pi$. The periodicity statement on the right extends it to every other time: any input outside the window is first slid by whole periods until it lands inside. Together they define $h$ for all $t$.

The plug-in procedure, as a machine

The three plug-ins we actually did, run through the machine

• 1$h(\pi /2)$: already inside the window (1.57 is between 0 and 6.28). Which case? 1.57 < 3.14, first line. Output 1.
• 2$h(-\pi /2)$: outside the window (negative). Slide +2π: lands at $3\pi /2$ (4.71). Which case? Between 3.14 and 6.28, second line. Output 0.
• 3For practice, $h(7\pi /2)$ (≈ 11.0): outside (too big). Slide −2π: $3\pi /2$. Second line. Output 0.

So what did “put in our two numbers” mean in the even test?

The equation $h(-t)=h(t)$ is a CLAIM to be checked, not a formula to substitute into. Checking it at $t=\pi /2$ means: evaluate the left side with the lookup machine ($h(-\pi /2)=0$), evaluate the right side with the lookup machine ($h(\pi /2)=1$), and compare the two finished numbers: $0\ne 1$, claim false. Every “plug in” in the symmetry entries was this lookup, run once per side.

Perspective: plenty of engineering functions are lookup-defined rather than formula-defined, gear ratios per gear, tax bands, a thermostat’s on/off state. The square wave is just the simplest lookup function in your course, and the whole point of the Fourier series is to REBUILD it out of formula-functions (sines and cosines), because formulas are what calculus can differentiate, integrate and manipulate.

The middle is not compulsory, ANY time strictly inside a shelf works ($1$, $2$, $0.4$, all fine, see the “is the 2 arbitrary?” entry). The middle is just the safest spot. But $0$ and $\pi$ specifically are the two WORST choices, each broken for its own reason.

Reason 1: 0 and π are jump instants, there is no height to read there

Look at the brace: $h=1$ for $0<t<\pi$ and $h=0$ for $\pi <t<2\pi$. Both inequalities are STRICT ($<$, not $\le$), so neither line covers $t=0$ or $t=\pi$ at all. The function is literally undefined at the jumps. And you can see why on the graph: approach $\pi$ from the left and the wave sits at 1; approach from the right and it sits at 0. Two different answers collide at the same instant, so “the height at π” has no single value. A symmetry test needs definite heights to compare; a jump cannot supply one.

(Extra wrinkle: the Fourier series itself, evaluated at a jump, converges to the MIDPOINT ½, a third value different from both shelves. Jumps are genuinely ambiguous territory, stay off them.)

Reason 2: t = 0 is useless for a symmetry test even if it were defined

The test compares the height at $t$ with the height at $-t$. Put $t=0$ in and the mirror partner is $-0=0$: the point is its own partner. The even test becomes “is $h(0)=h(0)$?”, which is true for EVERY function ever. It can never expose an asymmetry, so it carries zero information. A useful test point must be strictly away from zero so that $t$ and $-t$ are genuinely different places.

Reason 3: π’s mirror partner is ALSO a jump

Suppose you shrugged off Reason 1 and tried $t=\pi$ anyway. The test then needs the height at $-\pi$, and by periodicity $-\pi$ sits exactly where $+\pi$ does in the repeating pattern: on ANOTHER seam between a 1-shelf and a 0-shelf. Both ends of the comparison are ambiguous. Nothing comparable on either side.

The rule to carry

• 1Pick test times STRICTLY INSIDE a flat stretch, never at a jump, never at $t=0$.
• 2Check where the MIRROR PARTNER lands too: it must also sit inside a shelf, not on a seam.
• 3“Middle of the shelf” is a habit, not a law: it keeps you (and your partner point) as far as possible from every edge, so there is never doubt about which shelf you are reading.

Companion to the graph-building entry below: same wave ($h=1$ on $(0,\pi )$, $0$ on $(\pi ,2\pi )$, repeating every $2\pi$), and now the question “is it odd, even, or neither?” answered with nothing assumed.

Step 0: what the two symmetries even are

“Even” and “odd” are names for two ways a graph can be its own twin:

• EEVEN = mirror twin. Fold the picture along the vertical axis (the line at $t=0$). If the left half lands exactly on the right half, the function is even. In symbols: the height at $-t$ always equals the height at $t$, written $h(-t)=h(t)$.
• OODD = upside-down twin. Spin the whole picture half a turn (180°) about the origin point. If it lands on itself, the function is odd. In symbols: the height at $-t$ is always the NEGATIVE of the height at $t$, written $h(-t)=-h(t)$.

A function might have the first symmetry, the second, or NEITHER. To find out which, you compare heights at mirror-partner times: some time $t$ on the right of zero, and its partner $-t$ the same distance to the left. One pair of heights is enough to kill a symmetry, because a symmetry must hold everywhere; one place where it fails means it is gone.

Step 1: pick a test time and read the right-side height

Pick $t=\frac{\pi }{2}$ (≈ 1.57), a convenient spot in the middle of the ON shelf. From the graph (or the brace: 1.57 is between 0 and 3.14, the “height 1” stretch):

Step 2: read the left-side height, slowly (this is the step people skip)

Now we need the height at the mirror partner, $t=-\frac{\pi }{2}$ (≈ −1.57). Problem: the brace only describes times between $0$ and $2\pi$, and −1.57 is outside that window. The way in is periodicity: the wave repeats every $2\pi$, meaning sliding any time forward by one full period lands on an identical part of the wave. So slide −π/2 forward by $2\pi$:

(The arithmetic: $-\frac{\pi }{2}+2\pi =-\frac{\pi }{2}+\frac{4\pi }{2}=\frac{3\pi }{2}$, about 4.71.) And 4.71 lies between $\pi$ (3.14) and $2\pi$ (6.28), the OFF shelf, height 0:

Step 3: run the EVEN test

Even demands mirror partners have EQUAL heights: $h(-t)=h(t)$. Put in our two numbers:

The left dot is on the floor, the right dot is up at 1. Fold the picture along the vertical axis and those two points do NOT land on each other. Not even.

Step 4: run the ODD test

Odd demands mirror partners have OPPOSITE heights: $h(-t)=-h(t)$. Since $h(\pi /2)=1$, oddness would need the partner to be at $-1$:

There is also a zero-effort way to see oddness was never possible: an odd graph must dip BELOW the axis somewhere (every up-height needs a matching down-height after the 180° spin). This wave lives entirely at heights 0 and 1, never negative, so it cannot be odd no matter which points you test. Not odd.

Step 5: the verdict, and what it changes

• 1Even test failed, odd test failed → the wave is NEITHER. That one word is what the exam wants stated (with a reason, e.g. the failing pair of heights).
• 2Consequence for the series: no symmetry shortcut applies directly, so in principle all three coefficient families ($a_0$, $a_n$, $b_n$) must be computed from the brace.
• 3Hidden bonus (worked in the DC-term entry below): subtract the mean ½ and what remains IS odd, which is the deep reason the $a_n$ all come out zero anyway and the final series is a constant plus sines.

One caution worth engraving: a symmetry test that PASSES at one point proves nothing (symmetry must hold at every point), but a test that FAILS at one point is final. That is why one carefully chosen pair of heights settles “not even” and “not odd” here, yet no number of passing points could ever prove a symmetry on its own.

They are the same quantity, not merely related. In mechanics, angular velocity is how fast something turns: radians of angle per second. The Fourier $\omega$ is the angular velocity of an invisible rotating point whose shadow draws the wave. Same symbol, same formula, same units, on purpose.

The mechanics version first

A wheel turning steadily sweeps $2\pi$ radians (one full turn) every revolution. If one revolution takes $T$ seconds, the turning rate is:

That IS the formula from the tutorial. Nothing about it is special to waves yet, it is just “how fast is the turning”.

The bridge: a sine wave is circular motion seen edge-on

Take a point moving round a circle of radius $A$ at a steady $\omega$ rad/s. After $t$ seconds it has turned through the angle $\omega t$. Now watch only its height (its shadow on a vertical wall): that shadow is exactly $A\sin (\omega t)$. Every sinusoid is a rotation viewed from the side:

One full lap is $2\pi$ radians of turning AND one full period $T$ of the wave, which is exactly why $\omega =2\pi /T$ for waves too. The wave inherits its $\omega$ from the circle that generates it.

So what does ω = 1 rad/s mean for the square wave?

• 1The square wave itself does not rotate. But its Fourier series is built from harmonics $\sin (nt)$, $\cos (nt)$, and EACH of those is the shadow of its own spinning point.
• 2$\omega =1$ rad/s says the fundamental’s circle turns 1 radian per second, so a full lap ($2\pi$ rad) takes $2\pi$ seconds, matching the wave’s period $T=2\pi$. Consistent.
• 3The $n$-th harmonic’s circle spins $n$ times faster ($n\omega$ rad/s), completing $n$ laps per wave period. That is what “harmonics at multiples of the fundamental” means physically.

Why engineers keep ω around (and not just ν in Hz)

Frequency $\nu =1/T$ counts full turns per second (Hz); $\omega =2\pi \nu$ counts radians per second. Maths prefers $\omega$ because the argument of sin/cos must be an angle, writing $\sin (\omega t)$ keeps the bookkeeping in radians with no stray $2\pi$ factors. And the rotation picture is not a metaphor you discard later: it is literally the phasor used for AC circuits in PE2/AE2 (UK mains: $\nu =50$ Hz means a phasor spinning at $\omega =2\pi ·50\approx 314$ rad/s), and the same $\omega$ is a motor shaft’s angular velocity in dynamics. One concept, whole degree.

Goal: start from a blank page and end at the app’s graph, with no step assumed. The function is the one from Q2(a):

Step 0: what a graph even is

A graph is a lookup table drawn as a picture. The horizontal axis is the floor: every position on it is a time $t$. The vertical axis is the wall: every position on it is a height. A point drawn at floor-position $t$ and wall-height $y$ is the statement “at time $t$, the function’s value is $y$”. Drawing the whole graph means doing that for every time at once.

Before drawing anything, mark the landmark times the brace mentions. $\pi$ is just the number 3.14…, so the landmarks are $0$, $\pi \approx 3.14$, and $2\pi \approx 6.28$:

Step 1: draw the first line of the brace

Line one says: for every time between $0$ and $\pi$, the height is $1$. Pick any such time and plot its point: at $t=1$, height 1. At $t=2$, height 1. At $t=0.5$, height 1. Every single point lands at the SAME height, so joining them gives a flat shelf at height 1 running from $t=0$ to $t=\pi$:

Step 2: draw the second line, height 0 lies ON the axis

Line two says: for every time between $\pi$ and $2\pi$, the height is $0$. Height zero is not “nothing drawn”, it is a real shelf that happens to sit at the level of the floor itself. So the graph between $\pi$ and $2\pi$ is a flat segment lying exactly ON the t-axis. This is why the app’s picture looks like the wave “disappears” there: it has not disappeared, it is at height 0.

At $t=\pi$ itself, the value jumps: just left of $\pi$ the height is 1, just right of it the height is 0. Nothing is drawn vertically at the jump, a function has exactly one value per time, so there is no wall connecting the shelves. The dashed vertical in the app is only a visual guide marking where the teleport happens. (Fine print: the brace uses strict < signs, so at exactly $t=\pi$ it does not define a value at all; for Fourier purposes the series will later converge to the midpoint ½ there.)

The window we have drawn runs from 0 to $2\pi$, length $2\pi$. That length is the period: $T=2\pi$, hence $\omega =\frac{2\pi }{T}=1$ rad/s.

Step 3: photocopy the block both ways forever

“Periodic” means $h(t+2\pi )=h(t)$: shift the picture by one period and it must look identical. So stamp the Step-2 block at every shift of $2\pi$. Shift right: shelf at 1 on $(2\pi ,3\pi )$, zero on $(3\pi ,4\pi )$. Shift left: shelf at 1 on $(-2\pi ,-\pi )$, zero on $(-\pi ,0)$:

Step 4: verify against the app’s picture

• 1Just right of 0: shelf at 1, width π. Matches (the app’s hump from 0 to π). ✓
• 2Just LEFT of 0: the app shows the wave on the axis from −π to 0. Our copy rule predicts exactly that: $(-\pi ,0)$ is the zero shelf of the left-shifted block. ✓
• 3Spot values: $h(\pi /2)=1$ (inside the first hump), $h(3\pi /2)=0$ (on the zero shelf), matching the earlier reading-a-value entry. ✓

Contrast with Q2(b) below: same recipe exactly (draw the brace once, tile by the period), only the ingredients differ, Q2(b)’s shelves are −1 and +1 with unequal widths and the stamp distance is 3 instead of 2π. Master the recipe once and every “sketch over a few periods” question is the same two moves.

Yes, almost all of them, because the series is computed from the BRACE (the piecewise definition), not from your drawing. But there is exactly one pipe connecting the two, and if you use it carelessly a bad graph can poison the algebra. Know where that pipe is.

What each part is actually marked on

• 1The sketch marks are their own little pot: axes, correct widths and heights, a few periods shown. Lose these and they are gone, but they are only the sketch’s pot.
• 2The series marks come from: correct $T$ and $\omega =2\pi /T$, correctly split integrals with the right limits and the right constant in each piece, correct integration, correct assembly. Every one of those reads straight off the brace: for Q2(b), $b_n=\frac{2}{3}[{\int }_{-2}^0(-1)\sin (n\omega t)dt+{\int }_0^1(+1)\sin (n\omega t)dt]$. No picture required anywhere in that.

The one way a bad graph poisons the series

The symmetry shortcut. If your (wrong) sketch convinces you the wave is odd, you will write “odd, so $a_0=a_n=0$” and skip two integrals. If the wave is actually “neither” (as in Q2(b)), you have now thrown away real, non-zero coefficients, and the final series is wrong for a reason that traces straight back to the graph.

The safe play when unsure of your sketch

• ✓Don’t claim a symmetry you can’t verify from the brace. The algebraic test (compare h(−t) with ±h(t) using the piecewise rule, not the picture) is sketch-proof.
• ✓If in any doubt, compute all three coefficients from the brace. Costs a few minutes, immune to drawing errors, and every correct integral earns its marks even if the sketch above it is nonsense.
• ✓Use the graph only for free CHECKS, never as an input: the DC term should match the eyeballed average, and coefficients should decay like 1/n for a jumpy wave. If they disagree, something is wrong somewhere, which is information, not marks lost.

Bottom line for Q2(b): a mangled sketch costs you the sketch’s own couple of marks and nothing else, PROVIDED your series work starts from the brace and you only state “neither odd nor even” if the algebra (or the widths in the brace: 1 vs 2) backs it.

The question hands you a piecewise rule for ONE period and expects you to turn it into a repeating graph. There are exactly two steps: (1) draw literally what the brace says, once; (2) photocopy that block left and right forever. Here is each step slowly.

Step 1: read the brace, draw one block

The rule is:

Read each line as “flat shelf at this height, over this stretch of floor”: a shelf at height $-1$ from $t=-2$ to $t=0$ (width 2), then a shelf at height $+1$ from $t=0$ to $t=1$ (width 1). Together they cover $-2<t<1$, a window of total length $3$. That length IS the period: $T=1-(-2)=3$, so $\omega =\frac{2\pi }{3}$ rad/s.

Step 2: photocopy the block every 3 units

“Periodic with period 3” means $h(t+3)=h(t)$: slide the whole picture 3 to the right (or left) and it must look identical. So take the block from Step 1 and stamp copies at every shift of 3:

Where each feature of the final graph comes from:

• 1Narrow tops, wide bottoms: straight from the brace, the $+1$ piece is width 1, the $-1$ piece is width 2. (A one-third duty cycle.)
• 2Jump locations: the block’s internal jump at $t=0$ and its seams at $-2$ and $1$, then all of these repeated every 3: …, −2, 0, 1, 3, 4, 6, 7, …
• 3Dashed verticals: not part of the function, just guides marking the instant the value teleports between −1 and +1. A function can only have one value per t, so the jump has no “vertical wall”.

Why the graph shows “neither odd nor even” at a glance

The widths give it away. Mirror the graph in the vertical axis: just right of 0 the wave is $+1$ for a width-1 stretch, but just left of 0 it is $-1$, so it is not even. For oddness, beware: near the origin it LOOKS odd (heights flip across 0). Test one point further out: $h(1.5)=-1$ (in the (1,3) low shelf) but $-h(-1.5)=-(-1)=+1$. Not equal, so not odd. The unequal widths (1 vs 2) are what break both symmetries.

Lesson inside the lesson: a symmetry test that PASSES at one point proves nothing, symmetry must hold everywhere, and this wave passes the odd test near 0 yet fails it at 1.5. One failing point disproves; one passing point does not prove.

Bonus: what the graph already tells you about the series

Mean value by areas: over one period, area $=(2\times (-1))+(1\times (+1))=-1$, divided by the width 3 gives the DC term $-\frac{1}{3}$. The wave spends more time low than high, so its average sits below zero, and with no symmetry, expect both $a_n$ and $b_n$ to survive in the series.

This is from the both-odd-and-even proof below: we reached $f(t)=-f(t)$ and then “added $f(t)$ to both sides” to get $2f(t)=0$. The confusion: it can look like we claimed $-f(t)+f(t)=f(t)$. We did not. The two sides get the same treatment but produce different results.

Watch each side separately

Adding the same thing to both sides of an equation keeps it true. Do it and simplify each side on its own:

• LLeft side: it was $f(t)$, we add $f(t)$, and two copies of the same thing is twice the thing: $2f(t)$. This is where the 2 comes from.
• RRight side: it was $-f(t)$, we add $f(t)$, and anything plus its own negative cancels to $0$. This is where the 0 comes from. (Never $f(t)$: that would be $-1+1=1$, which is false.)

Same move with a plain x

If the $f(t)$ clutter is the problem, rename it: let $x=f(t)$, one unknown number. The proof line is then just ordinary equation-solving:

Sanity check with a number: could $x=5$ satisfy $x=-x$? That would need 5 = −5, false. Could $x=0$? 0 = −0 is true. Zero is the only number equal to its own negative, which is exactly what the algebra concluded, and (applied at every t) why the only both-odd-and-even function is the zero function.

Yes, but only one function in all of mathematics pulls it off: the zero function, $f(t)=0$ for every $t$ (the flat line lying exactly on the horizontal axis). Nothing else qualifies. Here is why.

The proof: force both rules at once

Suppose some function $f$ is BOTH. Then it must obey both defining equations at the same time:

Both equations describe the SAME quantity $f(-t)$ on their left, so their right-hand sides must be equal to each other:

Now add $f(t)$ to both sides:

And this holds for every $t$. So the only way to be both is to be zero everywhere. The single number that is its own negative is $0$, and a function that must equal its own negative at every point has no choice but to be flat zero.

The picture: the one shape both symmetries allow

The flat line on the axis is the only graph that survives BOTH symmetry operations. Mirror it left-to-right, unchanged. Spin it 180° about the origin, still unchanged. Any bump anywhere would break one of the two.

The takeaway on the three labels

• 1“Odd” and “even” are NOT opposites. Most functions are neither (like our square wave), some are one, and exactly one, $f=0$, is both.
• 2So “both” is a real but trivial case. If a symmetry argument ever forces your function to be both odd and even, it is telling you the function is identically zero, sometimes a useful sanity check that a coefficient must vanish.

Our unit square wave is nowhere near this: it sits at 1 half the time, so it is not the zero function, and (as shown in the entries below) it is neither odd nor even.

“Odd” and “even” are two specific kinds of symmetry a graph can have. A function can be one, the other, or neither. To decide, you run two tests. If both fail, the verdict is “neither”, which is what happened here.

What the two symmetries look like

Before testing, picture what you are testing FOR. These are the only two special mirrors that matter:

In symbols, comparing the height at $t$ with the height at $-t$:

The two heights we already know

From reading the wave earlier, we have a value and its mirror-partner:

• +$h(\frac{\pi }{2})=1$  (that time is in the ON block)
• −$h(-\frac{\pi }{2})=0$  (its mirror lands in the OFF block)

To DISPROVE a symmetry you only need one place where it fails, so this single pair of heights is enough to settle both tests.

Test 1 — is it EVEN?

Even demands $h(-t)=h(t)$. Put our two heights in:

$0$ is not $1$, so the even equation is false. Not even.

Test 2 — is it ODD?

Odd demands $h(-t)=-h(t)$. That would need $h(-\pi /2)$ to equal $-1$:

$0$ is not $-1$, so the odd equation is false too. Not odd.

Verdict

Even test: failed. Odd test: failed. A function that is neither even nor odd is called, simply, neither. That is the conclusion.

Two ways to see it instantly, no test point needed

• 1Can’t be odd: an odd graph must dip below the axis (every positive bit needs a matching negative bit rotated through the origin). This wave only ever sits at $0$ or $1$, never negative, so odd is impossible on sight.
• 2Can’t be even: even means a left/right mirror about the vertical axis. The ON block sits to the RIGHT of $0$ with nothing mirroring it on the left (that side is OFF), so the picture isn’t mirror-symmetric.

Why we bothered checking

Symmetry is a shortcut: an even function needs only cosine terms, an odd function only sine terms, and either one kills half the coefficients before you integrate. “Neither” means no free lunch from symmetry directly, BUT this wave is secretly (odd wave) + (constant), so once you subtract its mean it becomes odd, and the series is a DC term plus sines only. That is why checking symmetry first, even when the answer is “neither”, still pays off.

Short answer: yes, that particular 2 is arbitrary. But this is worth pinning down properly, because a maths page is full of numbers and some are free choices while others are locked by the problem. Telling them apart stops a lot of confusion.

The 2 in π/2 was MY choice (a test point)

Remember where $\pi /2$ came from: to check whether the square wave was odd or even, I needed to pick some time and read the wave’s height there and at its mirror. I picked $\pi /2$. Nothing forced that. I just wanted a convenient spot sitting safely inside the ON block $(0,\pi )$, away from the jumps at the edges. Any time in that block would have done the job:

To prove it is free, redo the whole odd/even test with t = 1 instead:

• 11 is in the ON block (since $1<\pi \approx 3.14$), so $h(1)=1$.
• 2Its mirror: $h(-1)=h(-1+2\pi )=h(5.28)=0$ (OFF block).
• 3$1\ne 0$ and $1\ne -0$, so “neither odd nor even.” Same verdict as π/2 gave.

The conclusion did not depend on the number I picked, which is exactly what “arbitrary” means here.

But most 2’s in Fourier are NOT free, they are forced

Do not carry the “it’s just a number” lesson too far. The other 2’s floating around this topic are locked by the maths and cannot be changed:

• πThe 2 in $\omega =\frac{2\pi }{T}$: forced. It is there because one full turn of a circle is $2\pi$ radians. Change it and $\omega$ is simply wrong.
• ½The 2 in the DC term $\frac{a_0}{2}$ and the $\frac{2}{T}$ in every coefficient: forced. They fall out of the orthogonality integrals (a ${\sin }^2$ or ${\cos }^2$ averages to $T/2$ over a period). That $T/2$ is what the $\frac{2}{T}$ cancels.

The rule of thumb

• ?Ask: if I swapped this number, would the answer still be right?
• ✓Yes → it was a free CHOICE (a test point, a convenient limit of integration, a sample value). Pick whatever is easiest.
• ✗No → it is FORCED by a definition or an identity (a full circle is $2\pi$, orthogonality gives $T/2$). Do not touch it.

So: the 2 in the test point π/2 is a free choice, and swapping it for 1 or π/4 changes nothing. The 2 in ω = 2π/T and in a₀/2 is forced, and swapping it breaks the maths. Both are “a 2”, but they are doing completely different jobs.

No trig, no integration here. This is just: you are handed a time, and you have to read off the wave’s height at that time. We go one tiny step at a time.

First: what does “h(t)” even mean?

Think of $h$ as a machine. You feed it a time, it gives you back a height. That is all the notation $h(t)$ means: “the height of the wave at time $t$.”

So what is $h$ on its own? It is simply the name of the wave, the way a person is named Tom. $h$ by itself is not a number, it is the whole rule/shape. The letter is arbitrary (could be $f$, $g$, $v$); here it is $h$, handy for height. When you bolt a time onto it, $h(1.57)$, THAT is a number: the height the wave named $h$ reaches at that time.

Wait — π/2 is horizontal, 1 is vertical. How can they be equal?

They are not equal, and this is the single most important thing to get straight. $h(\pi /2)=1$ is NOT saying “π/2 is the same as 1.” They live on two different axes and never get compared:

• ↔π/2 is a horizontal thing: a time, a spot along the floor. It answers “how far along?”
• ↕1 is a vertical thing: a height, how tall the wave stands. It answers “how high?”

The machine $h$ is the bridge between the two: you hand it a floor-position (the time), and it hands back a height. So $h(\pi /2)=1$ reads: “at the floor-position π/2, the wave is standing 1 tall.” The left side is a height, that is why the answer is a height.

Analogy: a car’s speedometer. “At 3 o’clock the speed was 50 mph” does not mean 3 = 50. The 3 is a time, the 50 is a speed; the speedometer links them. Same here: the time π/2 and the height 1 are different kinds of thing, and $h$ is what links them.

Second: this machine’s rule is a light switch on a timer

Our wave’s rule has just two settings. Written out, it says:

• 1For any time between 0 and π  →  the height is 1 (switch ON).
• 2For any time between π and 2π  →  the height is 0 (switch OFF).

Here is that rule drawn on a timeline. The blue stretch is where the height is 1, the grey stretch is where it is 0:

Third: turn π/2 into a normal number

$\pi$ is just a number, about 3.14. So $\pi /2$ means 3.14 shared into 2, which is about 1.57. Nothing more mysterious than that:

So the question “what is $h(\pi /2)$?” is really just “what is the height at time 1.57?”

Fourth: find where 1.57 sits on the timeline

Put 1.57 on the same timeline. Is it in the blue (ON) stretch or the grey (OFF) stretch? The ON stretch runs 0 to 3.14, and 1.57 is smaller than 3.14, so it lands inside the blue stretch, roughly in the middle of it:

Fifth: read off the height for that stretch

The blue stretch has height 1 everywhere. Time 1.57 is in the blue stretch. So the height at 1.57 is 1. Drawn on the actual wave, you go up from 1.57 on the bottom axis until you hit the wave, and you hit it at height 1:

Do it once more, so it sticks: h(3π/2)

Same four steps. Turn it into a number: $3\pi /2$ is $3\times 3.14/2\approx 4.71$. Where does 4.71 sit? It is bigger than 3.14, so it is past π, in the grey OFF stretch (3.14 to 6.28). The grey stretch has height 0. So:

That is the exact value used for h(−π/2) in the odd/even test: −π/2 is the same spot as 3π/2 once you slide over by one full period, and both read 0.

The whole thing in one sentence

• →Turn the time into a plain number.
• →See which stretch (ON or OFF) that number falls in.
• →Read off that stretch’s height. Done.

There is genuinely no calculation, it is a lookup. The wave only ever sits at 1 or 0, and which one you get depends only on whether your time landed before π or after it.

The setup: one period of the signal is $h(t)=1$ on $0<t<\pi$ and $h(t)=0$ on $\pi <t<2\pi$, repeated forever. A switch that is ON half the time, OFF half the time. Period $T=2\pi$, so $\omega =\frac{2\pi }{T}=1$ rad/s.

Part 1: odd, even, or neither, tested properly

The definitions are about comparing the function at $t$ and at $-t$:

Geometrically: even means the graph is a mirror image in the vertical axis; odd means the graph is unchanged by a 180° rotation about the origin. To DISPROVE a symmetry you only need one point where it fails, so the practical test is: pick a convenient $t$, read the height there and at $-t$, compare.

Take $t=\frac{\pi }{2}$. From the graph, $h(\frac{\pi }{2})=1$ (inside the ON block). For the height at $-\frac{\pi }{2}$, use periodicity, shifting by one whole period changes nothing:

because $\frac{3\pi }{2}$ lies in the OFF block $(\pi ,2\pi )$. Now run both tests at this one point:

• 1Even test: is $h(-t)=h(t)$?  Here $0\ne 1$. Fails. Not even.
• 2Odd test: is $h(-t)=-h(t)$?  Here $0\ne -1$. Fails. Not odd.

Both symmetries fail, so the wave is neither. There is also a faster argument against oddness that needs no point at all: an odd function must take the value $-h(t)$ whenever it takes $h(t)$, i.e. it needs matching NEGATIVE values. This wave only ever equals $0$ or $1$, it never goes below the axis, so it cannot possibly be odd.

The hidden symmetry: "neither" here means "odd plus an offset"

Here is the insight that explains the final answer before you compute anything. The wave itself has no symmetry, but subtract its average height $\frac{1}{2}$ and look at what is left:

$h(t)-\frac{1}{2}$ is a $\pm \frac{1}{2}$ square wave, and THAT function is genuinely odd. So the signal is really (odd function) + (constant). Odd functions expand in sines only, constants are the DC term, and that is the entire structure of the answer: a constant $\frac{1}{2}$ plus sines, with every $a_n$ destined to be zero. The calculation below just confirms it.

Part 2: what the DC term is

"DC" is borrowed from electronics: direct current, the steady, zero-frequency part of a signal, what a battery supplies, as opposed to the wiggling AC part. In a Fourier series the DC term is the constant out front, and it always equals the mean value of the signal over one period:

You can see the answer with no integration: the wave is at height $1$ for exactly half of every period and at $0$ for the other half, so its average height is $\frac{1}{2}$. That is the dashed line in the first figure, the level the wave "balances" around. Now the formula version, which is what the tutorial writes:

• 1The integral shrank from $(0,2\pi )$ to $(0,\pi )$ because the function is ZERO on the second half, integrating 0 adds nothing.
• 2Watch the factor of two: the course convention writes the constant as $\frac{a_0}{2}$, so $a_0=1$ but the DC level is $\frac{1}{2}$. Writing $1$ as the constant term is THE classic lost mark.
• 3Engineering check: this is a 50% duty-cycle switch on a 1 V rail, and its DC level is duty × amplitude = ½. The same idea runs PWM in Power Electronics: vary the duty cycle and you vary the DC you deliver.

Part 3: the remaining coefficients, briefly

The cosine coefficients die because $\sin (n\pi )=0$ at both limits (as the hidden-odd argument predicted):

The sine coefficients survive, split by the $\cos (n\pi )={(-1)}^n$ even/odd flip (see the entry below on exactly this):

Assemble constant + surviving sines:

Read the answer back against the structure we predicted: the $\frac{1}{2}$ is the DC term (the mean), and everything else is sines because the wave minus its mean is odd. Only odd harmonics appear (1st, 3rd, 5th…), the coefficients decay like $\frac{1}{n}$ because the wave has jumps, and at each jump the series converges to the midpoint $\frac{1}{2}$, which is, once again, the DC level.

The whole split is powered by one fact: $\cos (n\pi )$ is always either +1 or −1, flipping as $n$ steps through the integers. Plug those two values into $-\frac{\pi \cos (n\pi )}{n}$ and you get the two cases.

Step 1: cos(nπ) lands only on ±1

Walk the cosine graph at whole multiples of $\pi$. Every step of $\pi$ is half a wavelength, so you jump from one peak straight to the opposite one:

Even $n$ gives $+1$, odd $n$ gives $-1$. That is exactly what the compact identity says:

Step 2: substitute each value

Everything except the cosine, the $-\frac{\pi }{n}$, just rides along. Only the $\cos (n\pi )$ changes sign:

• E$n$ even, $\cos (n\pi )=+1$:  $-\frac{\pi }{n}·(+1)=-\frac{\pi }{n}$
• O$n$ odd, $\cos (n\pi )=-1$:  $-\frac{\pi }{n}·(-1)=+\frac{\pi }{n}$

The odd case is the one to watch: the two minus signs cancel (the leading $-$ times the $-1$ from the cosine), flipping the result positive. That sign flip is the entire content of the brace:

The one-line version

You do not have to write a brace at all. Because $\cos (n\pi )={(-1)}^n$, the same result folds into a single alternating term:

The bumped exponent, $n+1$ instead of $n$, is precisely that "minus signs cancel" flip written algebraically: $-{(-1)}^n={(-1)}^{n+1}$. Both forms are the same number; use the brace when a sketch or a physical sign matters, the ${(-1)}^{n+1}$ form when you are about to sum the series.

Where this shows up: it is the boundary term from $\int t\sin (nt)dt$ in the sawtooth $b_n$ coefficient. That is why the sawtooth series alternates in sign, $+,-,+,-$: the ${(-1)}^{n+1}$ is doing it.

Integration by parts is the product rule run backwards. It trades the integral you cannot do for one you can, and it is the tool behind every $\int t\sin (nt)dt$ and $\int t\cos (nt)dt$ in the Fourier unit.

The formula

With limits, the $uv$ term gets evaluated at them too:

Where it comes from: the product rule, reversed

Differentiate a product of two functions:

Integrate both sides with respect to $t$. The left side is a derivative being integrated, so it collapses back to $uv$:

Move one integral to the other side and you have the formula. Nothing deeper than that: by parts IS the product rule, read right-to-left.

How to pick u and dv

• 1$u$ = the part that gets simpler when differentiated. Powers of $t$ are the classic pick: $t$ differentiates to $1$ and vanishes from the leftover integral.
• 2$dv$ = the part you can integrate on sight: $\sin (nt)$, $\cos (nt)$, $e^{kt}$ all recycle forever under integration.
• 3Fourier habit: always $u=t$ (or $t^2$), $dv=$ the trig part. A $t^2$ needs by parts twice; each pass drops the power by one and pulls out a factor of $\frac{1}{n}$.

The workhorse, in full: ∫ t sin(nt) dt

Choose $u=t$ and $dv=\sin (nt)dt$. Then $du=dt$, and integrating $dv$ gives $v=-\frac{\cos (nt)}{n}$. Feed everything into the formula:

(The two minus signs on the leftover integral cancelled into a plus.) Finish with $\int \cos (nt)dt=\frac{\sin (nt)}{n}$:

The companion result, same moves with $dv=\cos (nt)dt$:

Where marks die: the minus sign in $v=-\frac{\cos (nt)}{n}$ and the double negative it creates one line later. Also: no +c is needed for $v$ itself, any one antiderivative works, the constant is added once at the very end (and not at all in definite integrals).

They come from the definition of polar coordinates. Polar describes a point not by "how far right, how far up" $(x,y)$, but by "how far from the origin, at what angle" $(r,\theta )$. The cos and sin are just the conversion between the two languages.

Step 1: the conversion formulas (SOH-CAH from the triangle)

Drop a vertical from the point down to the x-axis: you get a right triangle with hypotenuse $r$ and angle $\theta$ at the origin. The horizontal leg is $x$ (adjacent), the vertical leg is $y$ (opposite). Ordinary trigonometry:

Multiply each through by $r$ and you get the two substitution formulas:

Step 2: substitute both into x² + y²

(Squaring a product squares each factor: ${(r\cos \theta )}^2=r^2{\cos }^2\theta$.)

Step 3: factor out r² and use the Pythagorean identity

Both terms share $r^2$, so pull it out, and the bracket that remains is the most famous identity in trigonometry, equal to 1 for every angle:

Why ${\cos }^2\theta +{\sin }^2\theta =1$: it is Pythagoras on the same triangle. The legs are $r\cos \theta$ and $r\sin \theta$, the hypotenuse is $r$, so legs² sum to hypotenuse², and dividing through by $r^2$ leaves exactly that identity.

So the whole line in the notes is: substitute the polar definitions of $x$ and $y$, square them, factor, and the trig disappears, leaving the beautifully simple integrand $r^2$. That collapse is precisely why polar is the right tool whenever $x^2+y^2$ appears.

Because $25$ is $5^2$, and for a circle centred at the origin the number on the right of $x^2+y^2$ is the radius squared. So you read the radius off by square-rooting it.

Where $x^2+y^2=R^2$ comes from

A circle is every point at the same distance $R$ from the centre. The distance of a point $(x,y)$ from the origin is, by Pythagoras, $\sqrt{x^2+y^2}$. Set that equal to $R$ and square both sides:

So match it to 25

The region's boundary is $x^2+y^2=25$. Line that up with $x^2+y^2=R^2$:

(You take the positive root because a radius is a length.)

Why that instantly gives the polar limit

In polar, $r^2=x^2+y^2$, so the filled disk $x^2+y^2\le 25$ is just:

General rule: on a circle centred at the origin, $x^2+y^2=c$ is a circle of radius $\sqrt{c}$. The bare number is always the radius SQUARED, so square-root it. (Here $\sqrt{25}=5$; if it had been $9$ the radius would be $3$.)

Because once the inner integral is finished, $\frac{1}{3}{R}^3$ is a plain constant as far as $\theta$ is concerned: $R$ is a fixed radius, and there is no $\theta$ left in it. And a constant slides straight out of the integral.

The mechanics

Pull the constant to the front:

and the leftover integral is just integrate 1 across the interval, which measures the interval's width:

So the whole thing is the constant times $2\pi$:

The one-line picture

An integral is the area under a graph. Plot the constant height $\frac{1}{3}{R}^3$ against $\theta$ from $0$ to $2\pi$ and you get a flat line, so the region under it is simply a rectangle. No calculus needed once you see that.

Contrast: this shortcut works ONLY because $\theta$ dropped out of the integrand. If a $\theta$ had remained, say $\int \theta d\theta =\frac{{\theta }^2}{2}$, integrating would reshape it and you could not just multiply. "Contains no $\theta$" is the flag that the collapse to a single factor of $2\pi$ is allowed.